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lt 1228 1 1228fd typical application features description 100mhz current feedback amplifier with dc gain control the lt ? 1228 makes it easy to electronically control the gain of signals from dc to video frequencies. the lt1228 implements gain control with a transconductance ampli- fier (voltage to current) whose gain is proportional to an externally controlled current. a resistor is typically used to convert the output current to a voltage, which is then amplified with a current feedback amplifier. the lt1228 combines both amplifiers into an 8-pin package, and oper - ates on any supply voltage from 4v (2v) to 30v (15v). a complete differential input, gain controlled amplifier can be implemented with the lt1228 and just a few resistors. the lt1228 transconductance amplifier has a high imped- ance differential input and a current source output with wide output voltage compliance. the transconductance, g m , is set by the current that flows into pin 5, i set . the small signal g m is equal to ten times the value of i set and this relationship holds over several decades of set current. the voltage at pin 5 is two diode drops above the negative supply, pin 4. the lt1228 current feedback amplifier has very high input impedance and therefore it is an excellent buffer for the out- put of the transconductance amplifier . the current feedback amplifier maintains its wide bandwidth over a wide range of voltage gains making it easy to interface the transconduc- tance amplifier output to other circuitry . the current feed- back amplifier is designed to drive low impedance loads, such as cables, with excellent linearity at high frequencies. applications n very fast transconductance amplifier bandwidth: 75mhz g m = 10 i set low thd: 0.2% at 30mv rms input wide i set range: 1a to 1ma n very fast current feedback amplifier bandwidth: 100mhz slew rate: 1000v/s output drive current: 30ma differential gain: 0.04% differential phase: 0.1 high input impedance: 25m?, 6pf n wide supply range: 2v to 15v n inputs common mode to within 1.5v of supplies n outputs swing within 0.8v of supplies n supply current: 7ma n available in 8-lead pdip and so packages n video dc restore (clamp) circuits n video differential input amplifiers n video keyer/fader amplifiers n agc amplifiers n tunable filters n oscillators l, lt , lt c , lt m , linear technology and the linear logo are registered trademarks of linear technology corporation. all other trademarks are the property of their respective owners. lt1228 ? ta01 ? + ? + + + r3a 10k r2a 10k r3 100 r2 100 4.7f r4 1.24k r6 6.19k r5 10k i set r1 270 r g 10 r f 470 4.7f 15v g m cfa v out 1 8 6 3 2 7 5 4 ?15v + ? v in high input resistance even when power is off C18db < gain < 2db v in 3v rms frequency (hz) 100k ?24 gain (db) ?15 ?3 3 6 1m 10m 100m lt1228 ? ta02 0 ?6 ?9 ?12 ?18 ?21 i set = 100a v s = 15v r l = 100 i set = 1ma i set = 300a differential input variable gain amp frequency response
lt 1228 2 1228fd pin configuration absolute maximum ratings (note 1) order information lead free finish tape and reel part marking package description temperature range lt1228cn8#pbf lt1228cn8#trpbf lt1228cn8 8-lead plastic dip 0c to 70c lt1228in8#pbf lt1228in8#trpbf lt1228in8 8-lead plastic dip C40c to 85c lt1228cs8#pbf lt1228cs8#trpbf 1228 8-lead plastic so 0c to 70c lt1228is8#pbf lt1228is8#trpbf 1228i 8-lead plastic so C40c to 85c obsolete package lt1228mj8 lt1228mj8#trpbf lt1228mj 8 8-lead cerdip C55c to 125c lt1228cj8 lt1228cj8#trpbf lt1228cj8 8-lead cerdip 0c to 70c consult lt c marketing for parts specified with wider operating temperature ranges. consult lt c marketing for information on nonstandard lead based finish parts. for more information on lead free part marking, go to: http://www.linear.com/leadfree/ for more information on tape and reel specifications, go to: http:// www.linear.com/tapeandreel/ electrical characteristics the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25c. current feedback amplifier, pins 1, 6, 8. 5v v s 15v, i set = 0a, v cm = 0v unless otherwise noted. 8 7 6 5 4 3 2 1i out ?in +in v ? i set v out v + gain top view n8 package 8-lead pdip s8 package 8-lead plastic so + ? g m t jmax = 150c, ja = 100c/w (n) t jmax = 150c, ja = 150c/w (n) j8 package 8-lead cerdip t jmax = 175c, ja = 100c/w (j) obsolete package symbol parameter conditions min typ max units v os input offset voltage t a = 25c l 3 10 15 mv mv input offset voltage drift l 10 v/c i in + noninverting input current t a = 25c l 0.3 3 10 a a i in C inverting input current t a = 25c l 10 65 100 a a e n input noise voltage density f = 1khz, r f = 1k, r g = 10?, r s = 0? 6 nv/hz i n input noise current density f = 1khz, r f = 1k, r g = 10?, r s = 10k 1.4 pv/hz supply voltage ....................................................... 18 v input current , pins 1, 2, 3, 5, 8 ( note 8) .............. 15 ma output short circuit duration ( note 2) ......... continuous operating temperature range lt 1228 c ................................................... 0 c to 70 c lt 1228 i ................................................ C40 c to 85 c lt 1228 m ( obsolete ) ...................... C 55 c to 125 c storage temperature range .................. C65 c to 150 c junction temperature plastic package ................................................. 150 c ceramic package ( obsolete ) ......................... 175 c lead temperature ( soldering , 10 sec ) ................... 300 c
lt 1228 3 1228fd electrical characteristics the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25c. current feedback amplifier, pins 1, 6, 8. 5v v s 15v, i set = 0a, v cm = 0v unless otherwise noted. symbol parameter conditions min typ max units r in input resistance v in = 13v, v s = 15v v in = 3v, v s = 5v l l 2 2 25 25 m? m? c in input capacitance (note 3) v s = 5v 6 pf input voltage range v s = 15v, t a = 25c l 13 12 13.5 v v v s = 5v, t a = 25c l 3 2 3.5 v v cmrr common mode rejection ratio v s = 15v, v cm = 13v, t a = 25c v s = 15v, v cm = 12v v s = 5v, v cm = 3v, t a = 25c v s = 5v, v cm = 2v l l 55 55 55 55 69 69 db db db db inverting input current common mode rejection v s = 15v, v cm = 13v, t a = 25c v s = 15v, v cm = 12v v s = 5v, v cm = 3v, t a = 25c v s = 5v, v cm = 2v l l 2.5 2.5 10 10 10 10 a/v a/v a/v a/v psrr power supply rejection ratio v s = 2v to 15v, t a = 25c v s = 3v to 15v l 60 60 80 db db noninverting input current power supply rejection v s = 2v to 15v, t a = 25c v s = 3v to 15v l 10 50 50 na/v na/v inverting input current power supply rejection v s = 2v to 15v, t a = 25c v s = 3v to 15v l 0.1 5 5 a/v a/v a v large-signal voltage gain v s = 15v, v out = 10v, r load = 1k v s = 5v, v out = 2v, r load = 150? l l 55 55 65 65 db db r ol transresistance , ?v out /?i in C v s = 15v, v out = 10v, r load = 1k v s = 5v, v out = 2v, r load = 150? l l 100 100 200 200 k? k v out maximum output voltage swing v s = 15v, r load = 400?, t a = 25c l 12 10 13.5 v v v s = 5v, r load = 150?, t a = 25c l 3 2.5 3.7 v v i out maximum output current r load = 0?, t a = 25c l 30 25 65 125 125 ma ma i s supply current v out = 0v, i set = 0v l 6 11 ma sr slew rate (notes 4 and 6) t a = 25c 300 500 v/s sr slew rate v s = 15v, r f = 750?, r g = 750?, r l = 400? 3500 v/s t r rise time (notes 5 and 6) t a = 25c 10 20 ns bw small-signal bandwidth v s = 15v, r f = 750?, r g = 750?, r l = 100? 100 mhz t r small-signal rise time v s = 15v, r f = 750?, r g = 750?, r l = 100? 3.5 ns propagation delay v s = 15v, r f = 750?, r g = 750?, r l = 100? 3.5 ns small-signal overshoot v s = 15v, r f = 750?, r g = 750?, r l = 100? 15 % t s settling time 0.1%, v out = 10v, r f =1k, r g = 1k, r l =1k 45 ns differential gain (note 7) v s = 15v, r f = 750?, r g = 750?, r l = 1k 0.01 % differential phase (note 7) v s = 15v, r f = 750?, r g = 750?, r l = 1k 0.01 deg differential gain (note 7) v s = 15v, r f = 750?, r g = 750?, r l = 150 0.04 % differential phase (note 7) v s = 15v, r f = 750?, r g = 750?, r l = 150 0.1 deg
lt 1228 4 1228fd electrical characteristics the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25c. transconductance amplifier, pins 1, 2, 3, 5. 5v v s 15v, i set = 100a, v cm = 0v unless otherwise noted. symbol parameter conditions min typ max units v os input offset voltage i set = 1ma, t a = 25c l 0.5 5 10 mv mv input offset voltage drift l 10 v/c i os input offset current t a = 25c l 40 200 500 na na i b input bias current t a = 25c l 0.4 1 5 a a e n input noise voltage density f = 1khz 20 nv/hz r in input resistance-differential mode v in 30mv l 30 200 k? input resistance-common mode v s = 15v, v cm = 12v v s = 5v, v cm = 2v l l 50 50 1000 1000 m? m c in input capacitance 3 pf input voltage range v s = 15v, t a = 25c v s = 15v l 13 12 14 v v v s = 5v, t a = 25c v s = 5v l 3 2 4 v v cmrr common mode rejection ratio v s = 15v, v cm = 13v, t a = 25c v s = 15v, v cm = 12v l 60 60 100 db db v s = 5v, v cm = 3v, t a = 25c v s = 5v, v cm = 2v l 60 60 100 db db psrr power supply rejection ratio v s = 2v to 15v, t a = 25c v s = 3v to 15v l 60 60 100 db db g m transconductance i set = 100a, i out = 30a, t a = 25c 0.75 1.00 1.25 a/mv transconductance drift l C0.33 %/c i out maximum output current i set = 100a l 70 100 130 a i ol output leakage current i set = 0a (+i in of cfa ), t a = 25c l 0.3 3 10 a a v out maximum output voltage swing v s = 15v , r1 = v s = 5v , r1 = l l 13 3 14 4 v v r o output resistance v s = 15v, v out = 13v v s = 5v, v out = 3v l l 2 2 8 8 m m output capacitance (note 3) v s = 5v 6 pf i s supply current, both amps i set = 1ma l 9 15 ma thd total harmonic distortion v in = 30mv rms at 1khz, r 1 = 100k 0.2 % bw small-signal bandwidth r1 = 50?, i set = 500a 80 mhz t r small-signal rise time r1 = 50?, i set = 500a, 10% to 90% 5 ns propagation delay r1 = 50?, i set = 500a, 50% to 50% 5 ns note 1: stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. exposure to any absolute maximum rating condition for extended periods may affect device reliability and lifetime. note 2: a heat sink may be required depending on the power supply voltage. note 3: this is the total capacitance at pin 1. it includes the input capacitance of the current feedback amplifier and the output capacitance of the transconductance amplifier. note 4: slew rate is measured at 5v on a 10v output signal while operating on 15v supplies with r f = 1k, r g = 110? and r l = 400?. the slew rate is much higher when the input is overdriven , see the applications information section. note 5: rise time is measured from 10% to 90% on a 500mv output signal while operating on 15v supplies with r f = 1k, r g = 110? and r l = 100?. this condition is not the fastest possible, however, it does guarantee the internal capacitances are correct and it makes automatic testing practical. note 6: ac parameters are 100% tested on the ceramic and plastic dip packaged parts (j and n suffix) and are sample tested on every lot of the so packaged parts (s suffix). note 7: ntsc composite video with an output level of 2v. note 8: back to back 6v zener diodes are connected between pins 2 and 3 for esd protection.
lt 1228 5 1228fd typical performance characteristics total harmonic distortion vs input voltage spot output noise current vs frequency input common mode limit vs temperature small-signal control path bandwidth vs set current small-signal control path gain vs input voltage output saturation voltage vs temperature small-signal bandwidth vs set current small-signal transconductance and set current vs bias voltage small-signal transconductance vs dc input voltage transconductance amplifier, pins 1, 2, 3, 5 set current (a) 10 0.1 ?3db bandwidth (mhz) 1 10 100 100 1000 lt1228 ? tpc01 r1 = 100k r1 = 10k r1 = 1k r1 = 100 v s = 15v bias voltage, pin 5 to 4, (v) 0.01 transconductance (a/mv) 0.1 1 10 100 0.9 1.2 1.3 1.5 lt1228 ? tpc02 0.001 1.0 1.1 1.4 v s = 2v to 15v t a = 25c 1.0 10 100 1000 10000 0.1 set current (a) input voltage (mvdc) ?200 0 transconductance (a/mv) 0.2 0.4 1.4 2.0 ?150 ?100 ?50 200 lt1228 ? tpc03 0 100 150 1.8 1.6 1.2 0.6 0.8 ?55c v s = 2v to 15v i set = 100a 50 1.0 25c 125c input voltage (mv p?p ) 1 0.01 output distortion (%) 0.1 1 10 10 1000 lt1228 ? tpc04 i set = 100a v s = 15v i set = 1ma 100 frequency (hz) 10 10 spot noise (pa/hz) 100 1000 1k 100k lt1228 ? tpc05 v s = 2v to 15v t a = 25c 100 10k i set = 1ma i set = 100a temperature (c) ?50 v ? common mode range (v) 0.5 1.0 ?1.5 v + ?25 0 25 125 lt1228 ? tpc06 50 75 100 ?0.5 ?1.0 ?2.0 1.5 2.0 v ? = ?2v to ?15v v + = 2v to 15v set current (a) 10 1 ?3db bandwidth (mhz) 10 100 100 1000 lt1228 ? tpc07 v s = 2v to 15v v in = 200mv (pin 2 to 3) ?i out ?i set input voltage, pin 2 to 3, (mvdc) 0 0 control path gain (a/a) 1.0 120 200 lt1228 ? tpc08 ?i out ?i set 40 80 160 0.2 0.4 0.6 0.8 0.9 0.7 0.5 0.3 0.1 temperature (c) ?50 v ? output saturation voltage (v) +0.5 +1.0 ?1.0 v + ?25 0 25 125 lt1228 ? tpc09 50 75 100 ?0.5 2v v s 15v r1 =
lt 1228 6 1228fd typical performance characteristics voltage gain and phase vs frequency, gain = 20db C3db bandwidth vs supply voltage , gain = 10, r l = 100 C3db bandwidth vs supply voltage , gain = 10, r l = 1k voltage gain and phase vs frequency, gain = 40db C3db bandwidth vs supply voltage , gain = 100, r l = 100 C3db bandwidth vs supply voltage , gain = 100, r l = 1k voltage gain and phase vs frequency, gain = 6db C3db bandwidth vs supply voltage , gain = 2, r l = 100 C3db bandwidth vs supply voltage , gain = 2, r l = 1k transconductance amplifier, pins 1, 6, 8 frequency (mhz) 0 voltage gain (db) 2 4 6 8 0.1 10 100 lt1228 ? tpc10 ?2 1 7 5 3 1 ?1 phase shift (deg) 180 90 0 45 135 225 phase gain v s = 15v r l = 100 r f = 750 supply voltage (v) 2 ?3db bandwidth (mhz) 40 100 120 12 16 lt1228 ? tpc11 40 6 8 10 14 18 0 20 60 140 160 180 r f = 500 80 peaking 0.5db peaking 5db r f = 750 r f = 1k r f = 2k supply voltage (v) 2 ?3db bandwidth (mhz) 40 100 120 12 16 lt1228 ? tpc12 40 6 8 10 14 18 0 20 60 140 160 180 80 peaking 0.5db peaking 5db r f = 750 r f = 1k r f = 2k r f = 500 frequency (mhz) 14 voltage gain (db) 16 18 20 22 0.1 10 100 lt1228 ? tpc13 12 1 21 19 17 15 13 phase shift (deg) 180 90 0 45 135 225 phase gain v s = 15v r l = 100 r f = 750 supply voltage (v) 2 ?3db bandwidth (mhz) 40 100 120 12 16 lt1228 ? tpc14 40 6 8 10 14 18 0 20 60 140 160 180 r f = 500 80 peaking 0.5db peaking 5db r f = 750 r f = 1k r f = 2k r f = 250 supply voltage (v) 2 ?3db bandwidth (mhz) 40 100 120 12 16 lt1228 ? tpc15 40 6 8 10 14 18 0 20 60 140 160 180 r f = 500 80 peaking 0.5db peaking 5db r f = 750 r f = 1k r f = 2k r f = 250 frequency (mhz) 34 voltage gain (db) 36 38 40 42 0.1 10 100 lt1228 ? tpc16 32 1 41 39 37 35 33 phase shift (deg) 180 90 0 45 135 225 phase gain v s = 15v r l = 100 r f = 750 supply voltage (v) 2 ?3db bandwidth (mhz) 4 10 12 12 16 lt1228 ? tpc17 40 6 8 10 14 18 0 2 6 14 16 18 r f = 500 8 r f = 1k r f = 2k supply voltage (v) 2 ?3db bandwidth (mhz) 4 10 12 12 16 lt1228 ? tpc18 40 6 8 10 14 18 0 2 6 14 16 18 r f = 500 8 r f = 1k r f = 2k
lt 1228 7 1228fd typical performance characteristics input common mode limit vs temperature output saturation voltage vs temperature output short-circuit current vs temperature spot noise voltage and current vs frequency power supply rejection vs frequency output impedance vs frequency maximum capacitive load vs feedback resistor total harmonic distortion vs frequency 2nd and 3rd harmonic distortion vs frequency transconductance amplifier, pins 1, 6, 8 feedback resistor (k) 10 capacitive load (pf) 100 1k 10k 0 2 3 lt1228 ? tpc19 1 1 v s = 5v v s = 15v r l = 1k peaking 5db gain = 2 frequency (hz) total harmonic distortion (%) 0.01 0.10 10 1k 10k 100k lt1228 ? tpc20 0.001 100 v s = 15v r l = 400 r f = r g = 750 v o = 7v rms v o = 1v rms frequency (mhz) 1 ?70 distortion (dbc) ?60 ?50 ?40 ?30 ?20 10 100 lt1228 ? tpc21 v s = 15v v o = 2v p ? p r l = 100 r f = 750 a v = 10db 2nd 3rd temperature (c) common mode range (v) 2.0 v + ?50 25 75 125 lt1228 ? tpc22 v ? 0 1.0 ?1.0 ?2.0 ?0.5 ?1.5 1.5 0.5 ?25 50 100 v + = 2v to 15v v ? = ?2v to ?15v temperature (c) output saturation voltage (v) v + ?50 25 75 125 lt1228 ? tpc23 v ? 0 1.0 ?1.0 ?0.5 0.5 ?25 50 100 r l = 2v v s 15v temperature (c) ?25 output short-circuit current (ma) 40 60 100 150 lt1228 ? tpc24 0 ?50 25 50 75 125 175 30 70 50 frequency (hz) spot noise (nv/hz or pa/hz) 10 100 10 1k 10k 100k lt1228 ? tpc25 1 100 e n +i n ?i n frequency (hz) power supply rejection (db) 40 80 10k 1m 10m 100m lt1228 ? tpc26 0 100k v s = 15v r l = 100 r f = r g = 750 negative 20 60 positive frequency (hz) output impedance () 0.1 100 10k 1m 10m 100m lt1228 ? tpc27 0.001 100k 0.01 10 v s = 15v 1.0 r f = r g = 2k r f = r g = 750
lt 1228 8 1228fd typical performance characteristics simplified schematic current feedback amplifier, pins 1, 6, 8 setting time to 10mv vs output step setting time to 1mv vs output step supply current vs supply voltage settling time (ns) output step (v) 60 lt1228 ? tpc28 200 40 80 100 ?10 10 0 ?8 ?6 ?4 ?2 2 4 6 8 noninverting inverting v s = 15v r f = r g = 1k inverting noninverting settling time (s) output step (v) 12 lt1228 ? tpc29 40 8 16 20 ?10 10 0 ?8 ?6 ?4 ?2 2 4 6 8 noninverting inverting v s = 15v r f = r g = 1k noninverting inverting supply voltage (v) supply current (ma) 12 lt1228 ? tpc30 4 0 8 16 0 10 5 1 2 3 4 6 7 8 9 2 6 10 14 18 ?55c 25c 125c 175c lt1228 ? ta03 i out gain v out v + v ? Cin +in i set bias 1 2 3 5 6 4 7 8
lt 1228 9 1228fd applications information the lt1228 contains two amplifiers, a transconductance amplifier ( voltage - to- current ) and a current feedback ampli - fier (voltage - to- voltage ). the gain of the transconductance amplifier is proportional to the current that is externally programmed into pin 5. both amplifiers are designed to operate on almost any available supply voltage from 4v (2v) to 30v (15 v). the output of the transconductance amplifier is connected to the noninverting input of the current feedback amplifier so that both fit into an eight pin package. transconductance amplifier the lt1228 transconductance amplifier has a high imped- ance differential input (pins 2 and 3) and a current source output (pin 1) with wide output voltage compliance. the voltage to current gain or transconductance (g m ) is set by the current that flows into pin 5, i set . the voltage at pin ?5 is two forward biased diode drops above the nega- tive supply, pin 4. therefore the voltage at pin 5 (with respect to v C ) is about 1.2v and changes with the log of the set current (120mv/decade), see the characteristic curves . the temperature coefficient of this voltage is about C4mv/c (C3300ppm/c) and the temperature coefficient of the logging characteristic is 3300 ppm / c . it is important that the current into pin 5 be limited to less than 15ma. the lt1228 will be destroyed if pin 5 is shorted to ground or to the positive supply . a limiting resistor (2k or so) should be used to prevent more than 15ma from flowing into pin 5. the small- signal transconductance ( g m ) is given as g m = 10 ? i set , with g m in (a/v) and i set in (a).this rela - tionship holds over many decades of set current (see the characteristic curves ). the transconductance is inversely proportional to absolute temperature (C3300ppm/c ). the input stage of the transconductance amplifier has been designed to operate with much larger signals than is pos- sible with an ordinary diff-amp. the transconductance of the input stage varies much less than 1% for differential input signals over a 30 mv range (see the characteristic curve small - signal transconductance vs dc input voltage ). resistance controlled gain if the set current is to be set or varied with a resistor or potentiometer it is possible to use the negative temperature coefficient at pin 5 (with respect to pin 4) to compensate for the negative temperature coefficient of the transcon- ductance. the easiest way is to use an lt1004-2.5, a 2.5v reference diode, as shown below: temperature compensation of g m with a 2.5v reference lt1228 ? ta04 lt1004-2.5 v ? g m 5 4 r i set i set r v be v be 2.5v 2e g the current flowing into pin 5 has a positive temperature coefficient that cancels the negative coefficient of the transconductance. the following derivation shows why a 2.5 v reference results in zero gain change with temperature : since g m = q kt i set 3.87 = 10 ? i set and v be = e g ? akt q where a = in ct n ic ? ? ? ? ? ? 19.4 at 27 c c = 0.001, n = 3, ic = 100a ( ) e g is about 1.25v so the 2.5v reference is 2e g . solving the loop for the set current gives: i set = 2e g ? 2 e g ? akt q ? ? ? ? ? ? r or i set = 2akt rq
lt 1228 10 1228fd applications information substituting into the equation for transconductance gives : g m = a 1.94r = 10 r the temperature variation in the term a can be ignored since it is much less than that of the term t in the equa- tion for v be . using a 2.5v source this way will maintain the gain constant within 1% over the full temperature range of C55c to 125c. if the 2.5v source is off by 10%, the gain will vary only about 6% over the same temperature range. we can also temperature compensate the transconductance without using a 2.5 v reference if the negative power supply is regulated. a thevenin equivalent of 2.5v is generated from two resistors to replace the reference. the two resis- tors also determine the maximum set current, approxi- mately 1.1v/r th . by rearranging the thevenin equations to solve for r4 and r6 we get the following equations in terms of r th and the negative supply, v ee . r4 = r th 1? 2.5v v ee ? ? ? ? ? ? and r6 = r th v ee 2.5v temperature compensation of g m with a thevenin voltage diode drops above the negative supply, a single resistor from the control voltage source to pin 5 will suffice in many applications. the control voltage is referenced to the negative supply and has an offset of about 900mv. the conversion will be monotonic, but the linearity is determined by the change in the voltage at pin 5 (120mv per decade of current). the characteristic is very repeat- able since the voltage at pin 5 will vary less than 5% from part to part. the voltage at pin 5 also has a negative temperature coefficient as described in the previous sec- tion. when the gain of several lt1228s are to be varied together, the current can be split equally by using equal value resistors to each pin 5. for more accurate ( and linear ) control , a voltage - to- current converter circuit using one op amp can be used. the fol- lowing circuit has several advantages. the input no longer has to be referenced to the negative supply and the input can be either polarity (or differential). this circuit works on both single and split supplies since the input voltage and the pin 5 voltage are independent of each other. the temperature coefficient of the output current is set by r5. lt1228 ? ta05 r4 1.24k C15v g m 5 4 r' i set i set 1.03k v be v be v th = 2.5v r' r6 6.19k voltage controlled gain to use a voltage to control the gain of the transconductance amplifier requires converting the voltage into a current that flows into pin 5. because the voltage at pin 5 is two lt1228 ? ta19 r5 1k r1 1m v1 v2 i out to pin 5 of lt1228 50pf r1 = r2 r3 = r4 i out = ? = 1ma/v ? + r2 1m r3 1m r4 1m (v1 ? v2) r5 r3 r1 lt1006 digital control of the transconductance amplifier gain is done by converting the output of a dac to a current flow- ing into pin 5. unfortunately most current output dacs sink rather than source current and do not have output
lt 1228 11 1228fd applications information compliance compatible with pin 5 of the lt1228. there- fore, the easiest way to digitally control the set current is to use a voltage output dac and a voltage-to-current circuit . the previous voltage-to-current converter will take the output of any voltage output dac and drive pin 5 with a proportional current. the r, 2r cmos multiplying dacs operating in the voltage switching mode work well on both single and split supplies with the above circuit. logarithmic control is often easier to use than linear control. a simple circuit that doubles the set current for each additional volt of input is shown in the voltage controlled state variable filter application near the end of this data sheet. transconductance amplifier frequency response the bandwidth of the transconductance amplifier is a function of the set current as shown in the characteristic curves . at set currents below 100a, the bandwidth is approximately: C3 db bandwidth = 3 ? 10 11 i set the peak bandwidth is about 80mhz at 500a. when a resistor is used to convert the output current to a volt- age, the capacitance at the output forms a pole with the resistor. the best case output capacitance is about 5pf with 15v supplies and 6pf with 5v supplies. you must add any pc board or socket capacitance to these values to get the total output capacitance. when using a 1k resistor at the output of the transconductance amp, the output capacitance limits the bandwidth to about 25mhz. the output slew rate of the transconductance amplifier is the set current divided by the output capacitance, which is 6pf plus board and socket capacitance. for example with the set current at 1ma, the slew rate would be over 100v/s . transconductance amp small-signal response i set = 500a, r1 = 50? current feedback amplifier the lt1228 current feedback amplifier has very high noninverting input impedance and is therefore an excellent buffer for the output of the transconductance amplifier. the noninverting input is at pin 1, the inverting input at pin 8 and the output at pin 6. the current feedback ampli- fier maintains its wide bandwidth for almost all voltage gains making it easy to interface the output levels of the transconductance amplifier to other circuitry . the cur - rent feedback amplifier is designed to drive low imped- ance loads such as cables with excellent linearity at high frequencies. feedback resistor selection the small - signal bandwidth of the lt 1228 current feedback amplifier is set by the external feedback resistors and the internal junction capacitors. as a result, the bandwidth is a function of the supply voltage, the value of the feedback resistor, the closed-loop gain and load resistor. the char - acteristic curves of bandwidth versus supply voltage are done with a heavy load (100?) and a light load (1k) to
lt 1228 12 1228fd applications information show the effect of loading. these graphs also show the family of curves that result from various values of the feedback resistor. these curves use a solid line when the response has less than 0.5db of peaking and a dashed line for the response with 0.5db to 5db of peaking. the curves stop where the response has more than 5db of peaking. current feedback amp small -signal response v s = 15v, r f = r g = 750?, r l = 100? capacitance on the inverting input current feedback amplifiers want resistive feedback from the output to the inverting input for stable operation. take care to minimize the stray capacitance between the output and the inverting input. capacitance on the inverting input to ground will cause peaking in the frequency response (and overshoot in the transient response), but it does not degrade the stability of the amplifier. the amount of capacitance that is necessary to cause peaking is a func- tion of the closed-loop gain taken. the higher the gain, the more capacitance is required to cause peaking. for example, in a gain of 100 application, the bandwidth can be increased from 10mhz to 17mhz by adding a 2200pf capacitor, as shown below. c g must have very low series resistance, such as silver mica. at a gain of two, on 15v supplies with a 750? feedback resistor, the bandwidth into a light load is over 160mhz without peaking, but into a heavy load the bandwidth re- duces to 100mhz. the loading has so much effect because there is a mild resonance in the output stage that enhances the bandwidth at light loads but has its q reduced by the heavy load. this enhancement is only useful at low gain settings, at a gain of ten it does not boost the bandwidth. at unity gain, the enhancement is so effective the value of the feedback resistor has very little effect on the bandwidth . at very high closed-loop gains, the bandwidth is limited by the gain-bandwidth product of about 1ghz. the curves show that the bandwidth at a closed- loop gain of 100 is 10mhz, only one tenth what it is at a gain of two. lt1228 ? ta08 ? + c g r g 5.1 r f 510 v out cfa v in 6 1 8 boosting bandwidth of high gain amplifier with capacitance on inverting input frequency (mhz) 1 19 gain (db) 22 25 28 31 46 49 10 100 lt1228 ? ta09 34 37 40 43 c g = 4700pf c g = 2200pf c g = 0
lt 1228 13 1228fd applications information capacitive loads the lt 1228 current feedback amplifier can drive capacitive loads directly when the proper value of feedback resistor is used. the graph of maximum capacitive load vs feedback resistor should be used to select the appropriate value. the value shown is for 5db peaking when driving a 1k load, at a gain of 2. this is a worst case condition, the amplifier is more stable at higher gains , and driving heavier loads. alternatively, a small resistor (10? to 20?) can be put in series with the output to isolate the capacitive load from the amplifier output. this has the advantage that the amplifier bandwidth is only reduced when the capacitive load is present and the disadvantage that the gain is a function of the load resistance. slew rate the slew rate of the current feedback amplifier is not inde- pendent of the amplifier gain configuration the way it is in a traditional op amp. this is because the input stage and the output stage both have slew rate limitations. the input stage of the lt1228 current feedback amplifier slews at about 100v/s before it becomes nonlinear. faster input signals will turn on the normally reverse biased emitters on the input transistors and enhance the slew rate significantly . this enhanced slew rate can be as much as 3500v/s! current feedback amp large-signal response v s = 15v, r f = r g = 750? slew rate enhanced the output slew rate is set by the value of the feedback resistors and the internal capacitance. at a gain of ten with a 1k feedback resistor and 15v supplies, the output slew rate is typically 500 v/s and C850v/s. there is no input stage enhancement because of the high gain. larger feed- back resistors will reduce the slew rate as will lower supply voltages, similar to the way the bandwidth is reduced. current feedback amp large-signal response v s = 15v, r f = 1k, r g = 110?, r l = 400? settling time the characteristic curves show that the lt1228 current feedback amplifier settles to within 10mv of final value in 40ns to 55ns for any output step less than 10v. the curve of settling to 1mv of final value shows that there is a slower thermal contribution up to 20s. the thermal settling component comes from the output and the input stage. the output contributes just under 1mv /v of output change and the input contributes 300 v / v of input change . fortunately the input thermal tends to cancel the output thermal. for this reason the noninverting gain of two configuration settles faster than the inverting gain of one.
lt 1228 14 1228fd power supplies the lt1228 amplifiers will operate from single or split supplies from 2v (4v total) to 18v (36v total). it is not necessary to use equal value split supplies, however the offset voltage and inverting input bias current of the current feedback amplifier will degrade. the offset voltage changes about 350v/v of supply mismatch, the inverting bias current changes about 2.5a/v of supply mismatch. power dissipation the worst case amplifier power dissipation is the total of the quiescent current times the total power supply voltage plus the power in the ic due to the load. the quiescent supply current of the lt 1228 transconductance amplifier is equal to 3.5 times the set current at all temperatures. the quiescent supply current of the lt1228 current feedback amplifier has a strong negative temperature coefficient and at 150c is less than 7ma, typically only 4.5ma. the power in the ic due to the load is a function of the output voltage, the supply voltage and load resistance. the worst case occurs when the output voltage is at half supply, if it can go that far, or its maximum value if it cannot reach half supply. applications information typical applications for example, lets calculate the worst case power dis- sipation in a variable gain video cable driver operating on 12v supplies that delivers a maximum of 2v into 150?. the maximum set current is 1ma. p d = 2v s i smax + 3.5i set ( ) + v s ? v omax ( ) v omax r l p d = 2 ? 12v ? 7ma + 3.5 ? 1ma ( ) ? ? ? ? + 12v ? 2v ( ) 2v 150 ? = 0.252 + 0.133 = 0.385w the total power dissipation times the thermal resistance of the package gives the temperature rise of the die above ambient. the above example in so-8 surface mount pack- age (thermal resistance is 150c/w) gives: temperature rise = p d ja = 0.385w ? 150c/w = 57.75 c therefore the maximum junction temperature is 70c +57.75c or 127.75c, well under the absolute maximum junction temperature for plastic packages of 150c. basic gain control the basic gain controlled amplifier is shown on the front page of the data sheet. the gain is directly proportional to the set current. the signal passes through three stages from the input to the output. first the input signal is attenuated to match the dynamic range of the transconductance amplifier. the attenuator should reduce the signal down to less than 100mv peak. the characteristic curves can be used to estimate how much distortion there will be at maximum input signal. for single ended inputs eliminate r2a or r3a. the signal is then amplified by the transconductance amplifier (g m ) and referred to ground. the voltage gain of the transconductance amplifier is: g m ? r1 = 10 ? i set ? r1 lastly the signal is buffered and amplified by the current feedback amplifier ( cfa ). the voltage gain of the current feedback amplifier is: 1 + r f r g the overall gain of the gain controlled amplifier is the product of all three stages: a v = r3 r3 + r3a ? ? ? ? ? ? ? 10 ? i set ? r1 ? 1 + r f r g ? ? ? ? ? ? more than one output can be summed into r1 because the output of the transconductance amplifier is a current. this is the simplest way to make a video mixer.
lt 1228 15 1228fd typical applications video fader video dc restore (clamp) circuit lt1228 ? ta12 ? + ? + 1k 100 g m lt1223 cfa v out 3 2 5 1 v in1 1k ? + g m 3 1k v in2 10k 5.1k 10k 5.1k 10k 100 2 1k 5 1 ?5v v s = 5v the video fader uses the transconductance amplifiers from two lt1228s in the feedback loop of another cur - rent feedback amplifier, the lt1223. the amount of signal from each input at the output is set by the ratio of the set currents of the two lt1228s, not by their absolute value. the bandwidth of the current feedback amplifier is inversely proportional to the set current in this configuration. therefore, the set currents remain high over most of the pots range, keeping the bandwidth over 15mhz even when the signal is attenuated 20db. the pot is set up to completely turn off one lt1228 at each end of the rotation. lt1228 ? ta13 ? + ? + g m cfa v out 8 3 2 5 v ? 3k r f r g 4 10k 3k 5v logic input restore 0.01f video input 2n3906 1 6 v + 7 200 1000pf not necessary if the source resistance is less than 50 the video restore (clamp) circuit restores the black level of the composite video to zero volts at the beginning of every line. this is necessary because ac coupled video changes dc level as a function of the average brightness of the picture. dc restoration also rejects low frequency noise such as hum. the circuit has two inputs: composite video and a logic signal. the logic signal is high except during the back porch time right after the horizontal sync pulse. while the logic is high, the pnp is off and i set is zero. with i set equal to zero the feedback to pin 2 has no affect. the video input drives the noninverting input of the current feedback amplifier whose gain is set by r f and r g . when the logic signal is low, the pnp turns on and i set goes to about 1ma. then the transconductance amplifier charges the capacitor to force the output to match the voltage at pin 3, in this case zero volts. this circuit can be modified so that the video is dc coupled by operating the amplifier in an inverting configuration. just ground the video input shown and connect r g to the video input instead of to ground.
lt 1228 16 1228fd single supply wien bridge oscillator typical applications lt1228 ? ta15 ? + ? + g m cfa v o 8 3 2 1k 330 4 1 6 v + 7 1k 30pf 51 50 v o = 10db at v s = 5v all harmonics 40db down at v s = 12v all harmonics 50db down 9.1k 750 v ? 5 4.3k 4.7h 10k 0.1f v ? 2n3904 2n3906 in this application the lt1228 is biased for operation from a single supply. an artificial signal ground at half supply voltage is generated with two 10k resistors and bypassed with a capacitor. a capacitor is used in series with r g to set the dc gain of the current feedback amplifier to unity. the transconductance amplifier is used as a variable resistor to control gain. a variable resistor is formed by driving the inverting input and connecting the output back to it. the equivalent resistor value is the inverse of the gm. this works with the 1.8k resistor to make a variable attenuator. the 1mhz oscillation frequency is set by the wien bridge network made up of two 1000pf capacitors and two 160? resistors. for clean sine wave oscillation, the circuit needs a net gain of one around the loop. the current feedback amplifier has a gain of 34 to keep the voltage at the transconductance amplifier input low. the wien bridge has an attenuation of 3 at resonance; therefore the attenuation of the 1.8k resis- tor and the transconductance amplifier must be about 11, resulting in a set current of about 600a at oscillation. at start- up there is no set current and therefore no attenuation for a net gain of about 11 around the loop. as the output oscillation builds up it turns on the pnp transistor which generates the set current to regulate the output voltage. 12mhz negative resistance lc oscillator lt1228 ? ta14 ? + ? + g m cfa v o 8 3 2 5 r f 680 r g 20 4 1 6 v + 7 10k 10k 1.8k 160 1000pf 1000pf 160 + 10f + 10f v + 470 + 10f 100 0.1f 51 50 2n3906 6v to 30v f = 1mhz for 5v operation short out 100 resistor v o = 6dbm (450mv rms ) 2nd harmonic = C38dbc 3rd harmonic = ? 54dbc this oscillator uses the transconductance amplifier as a negative resistor to cause oscillation. a negative resistor results when the positive input of the transconductance amplifier is driven and the output is returned to it. in this example a voltage divider is used to lower the signal level at the positive input for less distortion. the negative resistor will not dc bias correctly unless the output of the transconductance amplifier drives a very low resistance. here it sees an inductor to ground so the gain at dc is zero. the oscillator needs negative resistance to start and that is provided by the 4.3k resistor to pin 5. as the output level rises it turns on the pnp transistor and in turn the npn which steals current from the transconductance amplifier bias input.
lt 1228 17 1228fd typical applications filters lt1228 ? ta16 ? + ? + g m cfa v out 8 3 2 r f 1k r2 120 5 1 6 r3 120 c 330pf f c = r3a 1k i set r g 1k r2a 1k v in lowpass input v in highpass input f c = 10 9 i set for the values shown 10 2 i set c r f + 1 r g r2 r2 + r2a single pole low/high/allpass filter allpass filter phase response frequency (hz) 10k phase shift (degrees) ?90 ?45 0 100k 1m 10m lt1228 ? ta17 ?135 ?180 1ma set current 100a set current using the variable transconductance of the lt1228 to make variable filters is easy and predictable. the most straight forward way is to make an integrator by putting a capacitor at the output of the transconductance amp and buffering it with the current feedback amplifier. because the input bias current of the current feedback amplifier must be supplied by the transconductance amplifier, the set current should not be operated below 10a. this limits the filters to about a 100:1 tuning range. the single pole circuit realizes a single pole filter with a corner frequency (f c ) proportional to the set current. the values shown give a 100khz corner frequency for 100a set current. the circuit has two inputs, a lowpass filter input and a highpass filter input. to make a lowpass filter, ground the highpass input and drive the lowpass input. conversely for a highpass filter, ground the lowpass input and drive the highpass input. if both inputs are driven, the result is an allpass filter or phase shifter. the allpass has flat amplitude response and 0 phase shift at low frequen- cies, going to C180 at high frequencies. the allpass filter has C90 phase shift at the corner frequency.
lt 1228 18 1228fd typical applications voltage controlled state variable filter lt1228 ? ta18 ? + ? + g m cfa lowpass output 8 3 2 1k 4 1 6 100 18pf ?5v 3.3k 7 5 ? + ? + g m cfa bandpass output 8 3 2 1k 4 1 6 100 18pf ?5v 3.3k 7 5 100 v in 5v 5v 3k 3k ? + 100pf lt1006 1k 180 10k v c 51k ?5v 2n3906 3.3k 3.3k f o = 100khz at v c = 0v f o = 200khz at v c = 1v f o = 400khz at v c = 2v f o = 800khz at v c = 3v f o = 1.6mhz at v c = 4v the state variable filter has both lowpass and bandpass outputs . each lt 1228 is configured as a variable integrator whose frequency is set by the attenuators, the capacitors and the set current. because the integrators have both positive and negative inputs, the additional op amp nor - mally required is not needed. the input attenuators set the circuit up to handle 3v pCp signals. the set current is generated with a simple circuit that gives logarithmic voltage to current control. the two pnp transistors should be a matched pair in the same package for best accuracy. if discrete transistors are used, the 51k resistor should be trimmed to give proper frequency response with v c equal zero. the circuit generates 100a for v c equal zero volts and doubles the current for every additional volt. the two 3k resistors divide the current between the two lt1228s. therefore the set current of each amplifier goes from 50a to 800a for a control voltage of 0v to 4v. the resulting filter is at 100khz for v c equal zero, and changes it one octave/v of control input.
lt 1228 19 1228fd package description please refer to http://www .linear.com/designtools/packaging/ for the most recent package drawings. j8 0801 .014 ? .026 (0.360 ? 0.660) .200 (5.080) max .015 ? .060 (0.381 ? 1.524) .125 3.175 min .100 (2.54) bsc .300 bsc (7.62 bsc) .008 ? .018 (0.203 ? 0.457) 0 ? 15 .005 (0.127) min .405 (10.287) max .220 ? .310 (5.588 ? 7.874) 1 2 3 4 8 7 6 5 .025 (0.635) rad typ .045 ? .068 (1.143 ? 1.650) full lead option .023 ? .045 (0.584 ? 1.143) half lead option corner leads option (4 plcs) .045 ? .065 (1.143 ? 1.651) note: lead dimensions apply to solder dip/plate or tin plate leads j8 package 3-lead cerdip (narrow .300 inch, hermetic) (reference ltc dwg # 05-08-1110) obsolete package
lt 1228 20 1228fd package description please refer to http://www .linear.com/designtools/packaging/ for the most recent package drawings. .016 ? .050 (0.406 ? 1.270) .010 ? .020 (0.254 ? 0.508) 45 0? 8 typ .008 ? .010 (0.203 ? 0.254) so8 rev g 0212 .053 ? .069 (1.346 ? 1.752) .014 ? .019 (0.355 ? 0.483) typ .004 ? .010 (0.101 ? 0.254) .050 (1.270) bsc 1 2 3 4 .150 ? .157 (3.810 ? 3.988) note 3 8 7 6 5 .189 ? .197 (4.801 ? 5.004) note 3 .228 ? .244 (5.791 ? 6.197) .245 min .160 .005 recommended solder pad layout .045 .005 .050 bsc .030 .005 typ inches (millimeters) note: 1. dimensions in 2. drawing not to scale 3. these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed .006" (0.15mm) 4. pin 1 can be bevel edge or a dimple s8 package 8-lead plastic small outline (narrow .150 inch) (reference ltc dwg # 05-08-1610 rev g) n8 rev i 0711 .065 (1.651) typ .045 ? .065 (1.143 ? 1.651) .130 .005 (3.302 0.127) .020 (0.508) min .018 .003 (0.457 0.076) .120 (3.048) min .008 ? .015 (0.203 ? 0.381) .300 ? .325 (7.620 ? 8.255) .325 +.035 ?.015 +0.889 ?0.381 8.255 ( ) 1 2 3 4 8 7 6 5 .255 .015* (6.477 0.381) .400* (10.160) max note: 1. dimensions are inches millimeters *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed .010 inch (0.254mm) .100 (2.54) bsc n package 8-lead pdip (narrow .300 inch) (reference ltc dwg # 05-08-1510 rev i)
lt 1228 21 1228fd information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no representa- tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. revision history rev date description page number d 06/12 updated order information table to new format clarified units used in g m = 10 ? i set relationship 2 9 (revision history begins at rev d)
lt 1228 22 1228fd linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 fax : (408) 434-0507 www.linear.com ? linear technology corporation 2012 lt 0612 rev d ? printed in usa related parts typical applications rf agc amplifier (leveling loop) + ? a3 lt1006 lt1228 ? ta20 + ? cfa 100 10k rf input 0.6v rms to 1.3v rms 25mhz 300 ?15v 15v + ? g m 470 10 0.01f 10k 0.01f 15v ?15v 4pf 10k 100k amplitude adjust 10k 4.7k ?15v lt1004 1.2v 10k output 2v p?p 1n4148?s couple thermally 3 2 7 5 1 8 4 inverting amplifier with dc output less than 5mv amplitude modulator lt1228 ? ta21 ? + + 100f ? + r5 v ? g m 3 2 v + 7 5 4 1 8 6 v o cfa r f 1k r g 1k v s = 5v, r5 = 3.6k v s = 15v, r5 = 13.6k v out must be less than 200mv p?p for low output offset bw = 30hz to 20mhz v in includes dc lt1228 ? ta22 ? + 1k ? + 10k ?5v g m 3 5v 7 5 4 1 8 6 v out 0dbm(230 mv) at modulation = 0v cfa r f 750 r g 750 + + 4.7f modulation input 8v p ? p 4.7f carrier input 30mv 2 part number description comments lt1227 140mhz current feedback amplifier 1100v/s slew rate, 0.01% differential gain, 0.03% differential phase LT1251/lt1256 40mhz video fader accurate linear gain control: 1% typ , 3% max lt1399 400mhz current feedback amplifier 800v/s slew rate, 80ma output current

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